1. Technical Field of the Invention
The present invention relates to a baseband gain control and particularly to a baseband gain control method and circuit capable of effectively preventing problems derived from a DC offset in the gain control of a direct conversion baseband circuit or the like.
2. Description of the Prior Art
A receiver utilizing direct conversion is advantageous over conventional super-heterodyne type receiver in the following respects and, therefore, expected to be widely used in the future:
1) A high frequency circuit section is simplified and the number of parts such as a filter can be reduced.
2) Since most of the functions including band limitation and AGC (automatic gain control) are executed at a baseband frequency, they can be realized by a CMOS analog circuit suited for LSI.
FIG. 6 is a view showing the concrete constitution of a direct conversion receiver. FIG. 6 shows a baseband gain control system for controlling the gain of a direct conversion baseband circuit, e.g., a system which has a wide dynamic range in the reception signals of a receiver of such a type as W-CDMA (Wide Band Code Division Multiple Access).
A high frequency signal received by an antenna 201 is subjected to band-limitation by a high frequency band-pass filter 202 and a received band is taken out. The signal thus band-limited is amplified by a low noise amplifier LNA 203 and directly inputted into a quadrature demodulator 204. The quadrature demodulator 204 is driven by a local signal generated by a local oscillator 225. The frequency of this local signal is the same as the central frequency of the received high frequency signal.
The quadrature demodulator 204 consists of multiplication circuits 222 and 223 and a phase circuit 224. The balanced outputs of the low noise amplifier LNA 203 are multiplied by the multiplication circuits 222 and 223 through an amplifier 221 in response to the balanced outputs of an orthogonal signal having a phase of 0xc2x0 and that of 90xc2x0 of the local signal, respectively, a baseband signal is directly generated from the high frequency signal, and two types of signals, i.e., baseband signals I and Q, are outputted as demodulated outputs. These baseband signals I and Q are subjected to band-limitation by baseband filters 205 and 206, respectively, and then amplified by an AGC circuit 207 so as to have a constant average amplitude.
The dynamic range of the AGC circuit 207 has characteristics of reaching several tens of decibels (about 80 dB for CDMA). The outputs of the AGC circuit 207 are outputted to the next stage as signals 215 and 216, respectively. It is noted that a circuit controlling the gain of this circuit and the algorithm thereof are unrelated to the present invention and, therefore, not described herein.
According to the direct conversion system, channel filters for suppressing adjacent channels are realized not by SAW filters for an IF band but by the baseband filters 205 and 206. Since they can be realized by circuits using active elements, the baseband filters 205 and 206 are suited for an IC. In addition, since the high frequency signal is directly converted into the baseband signals, there is no need to provide a second local oscillator. For these reasons, there is a probability that all the reception circuits from the low noise amplifier LNA 203 to the baseband outputs can be realized by one chip. This greatly contributes to making a cellular phone smaller in size and to the reduction of the number of parts.
Nevertheless, if there is a DC offset, even slightly, in the baseband filters 205 and 206 and the AGC circuit 207, the gain of the AGC sometimes becomes as high as 80 dB and a saturation phenomenon occurs that outputs are fixed to a power supply or the ground. For example, if there exists a DC offset of 1 mV in the bandpass filter 205 and the gain of the AGC circuit 207 is 80 dB, i.e., 10,000 times as high as an input, a DC component of 10 V is outputted. Needless to say, such a voltage is far beyond the voltage of a battery for a cellular phone, with the result that the cellular phone cannot operate.
As stated above, it is the most significant problem with the baseband circuit of the direct conversion circuit to eliminate a DC offset as much as possible.
There have been conventionally used high-pass filters (C-cut) each consisting of a DC cut capacitor or the like and provided between stages of variable gain amplifiers so as to eliminate the DC offset of a baseband circuit.
FIG. 7 is a view showing that the baseband circuit for I or Q shown in FIG. 6 is taken out. The baseband circuit consists of a plurality of gain control amplifiers having C-cut structures. To simplify description, FIG. 7 shows the baseband circuit as a single-end circuit. A baseband filter 101 and variable gain amplifiers 102, 103 and 104 (which amplifiers may be also referred to as xe2x80x9cVGA1xe2x80x9d, xe2x80x9cVGA2xe2x80x9d and xe2x80x9cVGA3xe2x80x9d, respectively) correspond to the baseband filter 205 (206) and the variable gain amplifiers 208 (211), 209 (212) and 210 (213), respectively.
According to this structure, for the purpose of preventing the propagation of a DC offset and the saturation of a signal due to the propagation thereof, high-pass filters 109 to 111 corresponding to C-cuts are inserted between the input section of the circuit and the VGA 102, the VGA 102 and VGA 103, the VGA 103 and the VGA 104 and the VGA 104 and the output section, respectively. The gains of the VGA1, VGA2 and VGA3 are controlled by gain control data distributed from the gain distribution circuit 112 based on gain data inputted from externally.
As stated above, by inserting the high-pass filters into the baseband circuit in appropriate units of the circuit, the propagation of a direct current is prevented in a static state in which gains have no change. In addition, the saturation of a signal due to the DC offset can be prevented.
However, according to the conventional method for eliminating a DC offset in the baseband circuit of the direct conversion receiver, a transient phenomenon due to the DC offset occurs in a dynamic control state in which gains have great change, which often has an adverse effect on reception characteristics.
Assuming that offset voltages Vof1, Vof2 and Vof3 are added to the input sides of the VGA1, VGA2 and VGA3, respectively, based on the circuit of FIG. 7, it is considered what type of a transient phenomenon occurs to an output if the respective gains g1, g2 and g3 are changed.
It is assumed here that the transfer functions of the high-pass filters 109 to 111 inserted as shown in FIG. 7 are the same and represented by the following expression for brevity.                               B          ⁡                      (            s            )                          =                              s                          s              +              α                                .                                    (        1        )            
It is assumed that the gains of the VGA1, VGA2 and VGA3 (not as dB values but as true values) are g1, g2 and g3, respectively, and that these gains are changed to g1xe2x80x2, g2xe2x80x2 and g3xe2x80x2, respectively. For brevity, the following conditions are set:
a) The gains g1, g2 and g3 are 1 time to 16 times as high as inputs;
b) The gains g1, g2 and g3 are not changed simultaneously; and
c) The gains g1, g2 and g3 are changed instantaneously.
1) If the gain of the VGA3 is changed from g3 to g3xe2x80x2:
Since being cut by the high-pass filters 109 and 110, respectively, the offset voltages Vof1 and Vof2 have no effect on the output and only the offset voltage Vof3 has an effect on the output. At the input of the high-pass filter 111, a step-like voltage change xcex94V3 occurs as follows.
xcex94V3=(g3xe2x80x2xe2x88x92g3)xc2x7vof3xe2x80x83xe2x80x83(2).
This step-like change influences an output Vout through the high-pass filter 111. A contribution thereof is described using Laplace transform as follows.                               V                      out            ⁡                          (              s              )                                      =                                            B              ⁡                              (                s                )                                      ·                                          Δ                ⁢                                  xe2x80x83                                ⁢                                  V                  3                                            s                                =                                    (                                                g                  3                  xe2x80x2                                -                                  g                  3                                            )                        ·                          V              of3                        ·                                          1                                  s                  +                  α                                            .                                                          (        3        )            
Assuming that g3 is changed at t=0, a time response is obtained as follows.
Vout(t)=(g3xe2x80x2xe2x88x92g3)xc2x7Vof3xc2x7exe2x88x92xcex1txe2x80x83xe2x80x83(4).
2) If the gain of the VGA2 is changed from g2 to g2xe2x80x2:
Because of the high-pass filter 110, the offset of the output of the VGA2 is cut by the filter 110 in a steady state. Then, it is assumed that g2 is changed to g2xe2x80x2. At this moment, the following step-like voltage change xcex94V2 occurs to the input of the high-pass filter 110.
xcex94V2=(g2xe2x80x2xe2x88x92g2)xc2x7Vof2xe2x80x83xe2x80x83(5).
This step-like change influences the output Vout through two stages of the high-pass filters. A contribution thereof is described using Laplace transform as follows.                               V                      out            ⁡                          (              s              )                                      =                                            g              3                        ·                                          B                ⁡                                  (                  s                  )                                            2                        ·                                          Δ                ⁢                                  xe2x80x83                                ⁢                                  V                  2                                            s                                =                                                    g                3                            ·              Δ                        ⁢                          xe2x80x83                        ⁢                                          V                2                            ·                              s                                  s                  +                  α                                            ·                                                1                                      s                    +                    α                                                  .                                                                        (        6        )            
Assuming that g2 is changed at t=0, a time response is obtained as follows:
Vout(t)=g3xc2x7xcex94V2xc2x7(1xe2x88x92xcex1xc2x7t)xc2x7exe2x88x92xcex1t=g3xc2x7(g2xe2x80x2xe2x88x92g2)xc2x7Vof2xc2x7(1xe2x88x92xcex1xc2x7t)xc2x7exe2x88x92xcex1txe2x80x83xe2x80x83(7).
3) If the gain of the VGA1 is changed from g1 to g1xe2x80x2:
Because of the high-pass filter 109, the offset of the output of the VGA1 is blocked by the filter 109 in a steady state. Then, it is assumed that g1 is changed to g1xe2x80x2. At this moment, the following step-like voltage change xcex94V1 occurs to the input of the high-pass filter 109.
xcex94V1=(g1xe2x80x2xe2x88x92g1)xc2x7Vof1xe2x80x83xe2x80x83(8).
This step-like change influences the output Vout through three stages of the high-pass filters. A contribution thereof is described using Laplace transform as follows.                               V                      out            ⁡                          (              s              )                                      =                                            g                              3                ⁢                                  xe2x80x83                                                      ·                          g              2                        ·                                          B                ⁡                                  (                  s                  )                                            3                        ·                                          Δ                ⁢                                  xe2x80x83                                ⁢                                  V                  1                                            s                                =                                                    g                3                            ·                              g                2                            ·              Δ                        ⁢                          xe2x80x83                        ⁢                                          V                1                            ·                              s                                  s                  +                  α                                            ·                              s                                  s                  +                  α                                            ·                                                1                                      s                    +                    α                                                  .                                                                        (        9        )            
Assuming that g1 is changed at t=0, a time response is obtained as follows.                                                                         V                                  out                  ⁡                                      (                    t                    )                                                              =                                                                    g                    3                                    ·                                      g                    2                                    ·                  Δ                                ⁢                                  xe2x80x83                                ⁢                                                      V                    1                                    ·                                      (                                          1                      -                                              2                        ·                        α                        ·                        t                                            +                                                                                                    α                            2                                                    ·                                                      t                            2                                                                          2                                                              )                                    ·                                      ⅇ                                                                  -                        α                                            ⁢                                              xe2x80x83                                            ⁢                      t                                                                                                                                              =                                                g                  3                                ·                                  g                  2                                ·                                  (                                                            g                      1                      xe2x80x2                                        -                                          g                      1                                                        )                                ·                                  V                  of1                                ·                                  (                                      1                    -                                          2                      ·                      α                      ·                      t                                        +                                                                                            α                          2                                                ·                                                  t                          2                                                                    2                                                        )                                ·                                                      ⅇ                                                                  -                        α                                            ⁢                                              xe2x80x83                                            ⁢                      t                                                        .                                                                                        (        10        )            
FIG. 8 shows the waveform of the mathematical expression (4) if the offset voltage Vof3 is 1 mV and the gain g3 is changed from 1 time to 16 times as high as the input.
FIG. 9 shows the waveform of the mathematical expression (7) if the offset voltage Vof2 is 1 mV and the gain g3 is 16 times as high as the input and the gain g2 is changed from 1 time to 16 times as high as the input.
FIG. 10 shows the waveform of the mathematical expression (10) if the offset voltage Vof1 is 1 mV, the gains g3 and g2 are 16 times as high as the inputs and the gain g1 is changed from 1 time to 16 times as high as the input.
In any case, the 3 dB-cutoff frequency of each high-pass filter is 5 kHz and the value of xcex1 is 31415.93.
As is obvious from FIGS. 8 to 10, even if a direct current component can be blocked by the high-pass filters, a high transient voltage occurs to the outputs and deteriorates characteristics by changing the gain of each stage inadvertently.
FIG. 8, for example, shows that if the gain g3 of the VGA3 is changed from 1 time (0 dB) to 16 times (24 dB) as high as the input with the DC offset voltage Vof3 of 1 mV, a transient voltage pulse of 1 mVxc3x97(16xe2x88x921)=15 mV occurs. FIG. 9 shows that if the gain g3 is 16 times as high as the input (24 dB) and the gain of the VGA2 is changed from 1 time (0 dB) to 16 times (24 dB) as high as the input with the DC offset voltage Vof2 of 1 mV, a transient voltage pulse of 1 mVxc3x9716xc3x97(16xe2x88x921)=240 mV occurs.
Further, FIG. 10 shows that if the gain g3 of the VGA3 is 16 times as high as the input (24 dB), the gain g2 of the VGA2 is 16 times as high as the input (24 dB) and the gain g1 of the VGA1 is changed from 1 time (0 dB) to 16 times (24 dB) as high as the input, a transient voltage pulse of 1 mVxc3x9716xc3x9716xc3x97(16xe2x88x921)=3840 mV occurs.
It is understood, therefore, that if the gains of a plurality of variable gain amplifiers are changed at random, a high transient voltage occurs to the output even with a low offset voltage. This transient voltage greatly damages the characteristics of the receiver.
As stated above, according to the direct conversion type receiver, it is necessary to control gains almost in a baseband frequency. Therefore, the saturation of amplifiers disadvantageously occurs due to the DC offset which occurs to the respective sections of the baseband circuit. To prevent this, a method for blocking the propagation of a DC component by providing high-pass filters at appropriate places of a circuit may be considered. In this case, however, a transient voltage occurs and deteriorates reception characteristics depending on gain change.
An object of the present invention is to provide a baseband gain control method and circuit capable of suppressing the generation of a transient voltage in the setting of the gains of a plurality of variable gain amplifiers in a baseband circuit.
The present invention is directed to control a baseband gain by setting gains of a plurality of variable gain amplifiers amplifying a baseband signal and connected to one another in series, to decrease the generation of a voltage due to a transient phenomenon. The settings of the gains of the plurality of variable gain amplifiers in the baseband circuit are controlled as follows:
1. A limitation is set to the quantity of the change of a gain which can be changed at one time. If a gain change exceeding the limiting value is necessary, the gain change is divided into a plurality of quantities of change each equal to or lower than the limiting value and a required gain change is attained while controlling the plurality of quantities of change a plurality of times.
2. If the gain is to be increased, the gains of the variable gain amplifiers starting at the variable gain amplifier close to an input are sequentially increased. If the gain is to be decreased, the gains of the variable gain amplifiers starting at the variable gain amplifier farthest to the input are sequentially decreased.
3. The generation of a transient voltage is suppressed by the gain control by the method of 1 or 2 or the combination of the methods 1 and 2.
According to the present invention, gain control is carried out so that the upper limit of the quantity of the change of the gains of a plurality of variable gain amplifiers for a baseband signal is set. Further, if the gain is to be increased, the gains of the variable gain amplifiers starting at the variable gain amplifier close to an input are sequentially increased. Further, if the gain is to be decreased, the gains of the variable gain amplifiers starting at the variable gain amplifier farthest to the input are sequentially decreased. Thus, it is possible to effectively suppress the generation of a transient voltage caused by a DC offset during the gain control.
The combination of the gain control for providing the upper limit of the quantity of the change of a gain and the distribution control for distributing different gains to a plurality of variable gain amplifiers according to the increase and decrease of the gain enables, in particular, suppressing the generation of a transient voltage caused by a DC offset more effectively.
If the present invention is applied to the gain control of a direct conversion baseband circuit, e.g., the baseband gain control of a direct conversion baseband circuit having a wide dynamic range of a reception signal such as a receiver of a W-CDMA (Wide Band Code Division Multiple Access) type, the present invention exhibits considerably great advantage.